Tissue signature tracking tranceiver

ABSTRACT

An ultrasound tranceiver providing enhanced imaging by selective filtering of the received signal to provide a variable frequency, constant bandwidth filtering of the received echo signals. The resulting signals are then detected to produce a signal, which when displayed, has a reduced number of false multiple images and enhanced signal quality from the deeper tissue discontinuities. Additionally, the received signal is detected in quadrature by reference to a simulated carrier pilot tone having a nonconstant frequency relationship to the transmitted signal. The resulting signal is post-processed to provide information which is used to display structural features, and in addition, the velocity profile of blood flow. The structure and velocity image information is superimposed to provide a composite signal wherein the static and dynamic characteristic of a patient is completely reported to the observer.

FIELD OF THE INVENTION

The present invention relates to medical ultrasound systems, and inparticular to ultrasound methods and apparatus for processing signals toprovide position and velocity information in real-time.

BACKGROUND OF THE INVENTION

For years, clinical ultrasound systems measurement has incorporatedeither pulsed or continuous wave (CW) ultrasound techniques for imagingof tissue structure and flow of blood therethrough. Since the tissueinvestigated is a dispersive medium, the signal transmitted into andthereafter reflected from tissue discontinuities suffers significantattenuation. That is, the greater the path taken by the acoustic signalwithin the subject, the greater the signal is attenuated and otherwisechanged. Previous systems have included compensation techniques such astime-controlled gain to provide correction for the anticipatedattenuation by the signal in the subject tissue. Similarly, othercorrection techniques have been applied with varying degrees of success.

Regardless of the correction techniques used thus far, certain problemsremain. Typical of these problems are the multiple reflections incurredbetween the specimen surface to be investigated and the surface and thetransducer within the ultrasonic probe. Moreover, for the deeper signalpenetration levels, the signal becomes uncorrectably attenuated andunfocussed, often obscuring critical imaging information.

These problems remain, since existing Doppler systems typically assume asignal propagation model incomplete for all applications. As a result,the resolution and/or range of Doppler or pulse-Doppler systems areunnecessarily limited.

SUMMARY OF THE INVENTION

The ultrasound tranceiver of the present invention provides avariable-frequency, fixed-bandwidth reception of the signals reflectedfrom tissue interfaces within the subject, providing improved rejectionof multiple surface reflections, combined with improved signalresolution at the extreme penetration depth.

The present invention further provides quadrature signal detection,which includes the recovery of phase information to provide a usefulsignal over a dynamic range of signals greater than previously known. Inparticular, useful information is provided even when the ultrasoundsignal drops well into the noise level of the receiver. Moreover, thesignals are sufficiently descriptive of the tissue structure to provideextreme position detail thereof. The newly added position detailinformation is used to determine the velocity profile of the blood flow.The system according to the present invention simultaneously providesthe tissue structure and provides blood flow velocity signals, whichwhen combined, provide an image having a composite of static and dynamictissue conditions.

Moreover, the present invention describes an ultrasound tranceiverhaving a detected signal, which inherently deconvolves the interactionof the transmitted pulse signal from the tissue reflection response,providing an enhanced signal resolution.

A further feature of the present invention is a combination ofultrasound tranceiver and transducer, wherein the ultrasound transducer,although apparently mistuned relative to the transmitted frequency,reduces the dependence on external circuitry for the time-controlledgain signal correction function.

BRIEF DESCRIPTION OF THE DRAWING

These and other features of the present invention can be betterunderstood by reading the following detailed description, taken togetherwith the drawing, wherein:

FIG. 1 is a pictorial representation of a typical ultrasound probe;

FIG. 2 is a waveform showing a typical excitation signal for thetransducer crystal;

FIG. 3 is a curve showing the frequency spectrum associated with thesignal of FIG. 2;

FIG. 4 is a curve showing a typical spectral distribution of thetransmitted wave;

FIG. 5 is a family of curves showing the transition and frequencydistribution relative to depth of signal reflection;

FIG. 6 is a sequence of waveforms showing the time domain signalcorresponding to FIG. 5;

FIG. 7 is an illustration showing multiple reflection signals;

FIG. 8 is a graph showing the relative decay rates of multiplereflection signals of FIG. 7;

FIG. 9 is a sequence of frequency curves demonstrating the variablebandpass concept according to the present invention;

FIG. 10 is a sequence of frequency curves showing the variable bandpassconcept combined with the time-gain compensation;

FIG. 11 is a sequence of waveforms of several signals of the presentinvention;

FIG. 12 is a collection of waveforms showing several signals related tothe transducer operation;

FIG. 13 is a collection of waveforms showing the effect of variousfilters on the transducer signals;

FIG. 14 is a drawing which illustrates a point spread function;

FIG. 15 is a block diagram of one embodiment of the movable passbandreceiver according to the present invention;

FIG. 16 is a collection of various waveforms of the system of FIG. 15;

FIG. 17 is a collection of curves showing the movable highpass andlowpass filter curves of the system of FIG. 15;

FIG. 18 is a collection of signal waveforms illustrating typical signaldefects;

FIG. 19 is a group of signal waveforms showing the alignment of a pulsesignal and a reference signal according to the present invention;

FIG. 20 is a collection of waveforms illustrating the sliding frequencyreference pilot signal;

FIG. 21 is a collection of curves showing a plot of the time-frequencymodulation of the reference pilot tone;

FIG. 22 is a collection of curves showing the motion of the spectralpeaks of acoustic reflection corresponding to the change in pilot toneshown in FIG. 21;

FIG. 23 is a group of time-related signal waveforms showing the productdetector output of a second embodiment of the present invention;

FIG. 24 is a phasor representation of the product detector outputsignals of a second embodiment of the present invention;

FIG. 25 is a phasor representation illustrating sine and cosine signalcomponents of a second embodiment according to the present invention;

FIG. 26 is a block diagram of a second embodiment of the ultrasoundtranceiver according to one embodiment of the present invention;

FIG. 27 is a block diagram of a quadrature signal post-processoraccording to one embodiment of the present invention;

FIG. 28 is an amplitude-versus-time waveform representation of thesignals received according to the present invention, distinguishingstatic features from blood flow velocity information;

FIG. 29 is a phasor diagram illustrating a low-clutter received signal;

FIG. 30 is a phasor diagram illustrating a high-clutter received echosignal; and

FIG. 31 is a phasor diagram illustrating the quadrature post-processoroperation wherein true velocity information is recovered from anotherwise high-clutter signal according to the system of FIGS. 27 and28.

DETAILED DESCRIPTION OF THE INVENTION

Many commercially available real-time scanners employ linear,trapezoidal, or sector scanning formats, wherein the vertical directionis proportional to echo time elapsed after the outgoing pulse, andproportional to depth in the patient (constant velocity-of-soundapproximation, currently a popular concept). Typical embodiments ofvarious known ultrasound systems are shown in several publications, suchas Doppler Ultrasound and Its Clinical Measurement, by Peter Atkinsonand John P. Woodcock, Academic Press, New York, 1982, incorporated byreference. Depending upon the style of the probe, any one of these threeformats can be scanned within the patient and displayed electronically.

A typical probe 20 suited for real-time diagnostic medical ultrasoundimaging is shown in FIG. 1. A crystal 4 emits an acoustical pulse 6along axis 8. This pulse first travels through an acousticallytransmissive liquid 10 and then through an acoustically transparentwindow 12. The pulse then passes through an acoustically transmissivecoupling jelly 14 and proceeds into the patient 16. Echoes from anatomyof interest, such as kidney 18 return along axis 8 through interfaces16, 12, 10 back to crystal 4. The crystal 4, acting also as amicrophone, generates feeble radio-frequency (RF) information containingamplitude, phase, and temporal information contributing to a single lineof imaging information. Drive means 24 cause the beam 8 to scan withinregion 26 at a rate sufficient to follow resporatory and cardiac-relatedpulsatile motion (e.g. 20 frames per second), but within the limitationthat the distance between adjacent lines is less than the azimuthalresolution of the crystal.

A commercially expedient method of exciting the crystal 4, to produce anoutgoing acoustical pulse, is to provide the crystal 4 with an"avalanche breakdown" voltage waveform, FIG. 2. When applied to acrystal 4, the waveform 30 causes the acoustically emitted pulse to havea shape shown in FIG. 3 due to bandpass limitations of a commercialcrystal of sufficient electromechanical efficiency, FIG. 3.

The frequency spectrum associated with the pressure waveform of FIG. 3is fairly broad.

FIG. 4 exemplifies a typical spectral distribution of such a transmittedwave 38, in the case of a general-purpose (abdominal) scanner. Signalsreflected from the body do not have the same spectrum, however, forseveral reasons. The most significant reason is that most of the humanbody attenuates sound at a rate proportional to frequency. This meansthat a pulse having many frequencies going into the body will not returnthese same frequencies in equal proportions. Shallow reflections willproduce spectral distributions and pulse shapes similar to the outgoingpulse but deeper reflections progressively distort spectra and pulseshapes so as to favor the lower frequency components.

FIG. 5 shows that deeper echoes become weaker and have less relativehigh-frequency components. FIG. 5 illustrates, typically, a single,specific reflection from the kidney (i.e., the same anatomical target,but placed at varying depths), showing the shallow reflection spectrum40 to be like the superficial reflection (at a 3-cm depth) of a kidneytransplant patient. FIG. 5 also shows the middle reflection 42 to belike the kidney image (at a 6-cm depth) of a very thin person and showsthe deep reflection 44 to be the same kidney feature (at 13-cm depth) inan obese patient. The purpose of this discussion is to illustrate thatthe acoustically detected signature of a known uniform (overall depth)anatomical feature changes with depth and that recognition of this facthas to be incorporated in the signal processing section of an ultrasoundscanner to produce an image having the least possible artificiallyinduced feature variance with depth.

FIG. 6 summarizes the same events in the time domain. The shallowreflection 50 has a waveform approximating the transmitted pulse 48. Themiddle reflection 52 and the deep reflection 54 show a progressivelylower "carrier frequency" with increasing depth, while preserving aroughly constant envelope shape and duration.

The combined interpretations of FIGS. 5 and 6 lead one to believe thatthe envelope characteristics are related to the percentage bandwidth(the "Q") of the crystal; however the sliding frequency characteristicsof echoes summarize the human body's "modulation" of an initially high"center ringing frequency" related to the fundamental frequency ofresonance of the crystal.

As in any communications system, the pulse-echo cycle involving thehuman body both as signal source (echo) and transmission medium must beoptimized in the "maximum channel capacity" sense. This conceptmaximizes the amount of retrievable imaging information. Specifically,both the two-dimensional spatial resolution and the dynamic range(signal-to-noise ratio) must be simultaneously optimized. Fixed focuscrystals come much closer to optimizing both resolution and dynamicrange simultaneously than do, for instance, phased-array transducers.Also, fixed-focus crystal scanners can be commercially implemented at alower cost, resulting in a higher image quality-to-dollar ratio.

Unfortunately, the price paid in fixed focus crystal systems is that amoving crystal is usually mounted away from the skin surface in a liquidfilled bath of 6 to 40 mm. A typical mechanical probe of the sectorscanning type (20 of FIG. 1) will employ an internal standoff of about 1cm. The presence of such a standoff (a practical necessity to keep theoscillating crystal from contacting the patient) causes a primaryacoustical reflection in the displayed image at a range of 1 cm, anddecreasing amplitude "multiple" reverberations at 2, 3, 4 cm, etc., assummarized in FIG. 7.

Such "multiple" echoes are due to the fact that the first reverberations(from the inside of the front cover) are not completely absorbed by thecrystal. The crystal absorbs only part of the first reflection; theremaining energy is re-emitted as though it were a second pulse, and thecycle repeats itself in an exponentially decreasing fashion. When theprobe contacts the patient, the first reflection from the front covergreatly diminishes due to the coupling to the patient, and the entiremultiple echo-train likewise diminishes; however, the diminishedmultiples can and frequently are present within the anatomical image atan amplitude (-20 to -30 dB) high enough to confuse diagnosticinterpretations.

However, the physics of multiples differ from the echo events in thepatient because of the liquid in the probe. Virtually every liquidcommonly used in mechanical probes has an acoustical absorptionproportional to the square of the frequency, whereas the acousticalabsorption in the patient is proportional to the first power offrequency. If one were to examine the multiple echogenicity at a highfrequency and then at a low frequency spectral region, one would observean entirely different rate of decay of the exponential series as shownin FIG. 8. By combining the notion that multiples can be suppressed bymaking the receiver insensitive to low frequencies for shallow depthswith the idea that the body tends to be frequency-selective on adescending scale vs depth, one can maximize the "patient information tomultiple clutter ratio" by operating the receiver at a fixed bandwidththat slides from a high passband to a low passband frequency mode duringeach pulse-echo cycle.

The sliding-frequency, fixed-bandwidth receiver mode can also beexplained in the context of a "transmission line." The effect of thetissue on the impulse response of the crystal in the body-machineinteraction, if viewed in the transmission line context, can be viewedas a classic case of envelope reconstruction; i.e., the information"pixel" packet can be approximated as a continuous wave (CW) carriermultiplied by either a Gaussian envelope or by a (sin kt/t^(n))²envelope. In practical "television" art terms, the events are comparableto receiving the same picture information (rise-time, axial resolution,etc.) on a channel carrier that progressively slides downhill. Thepractical implication to ultrasound is that a constant absolutebandwidth (having constant axial resolution) is consistent with therelatively constant envelope lengths 50, 52, 54 of FIG. 6, and that thedescending carrier frequency vs. depth due to tissue signaturemodulation is consistent with a downward sliding receiver retuning vs.echo time. By employing a movable frequency fixed bandwidth design,shown as filter passbands 60, 62, and 64 in sequence, of FIG. 9, oneobtains the optimum axial resolution vs. video noise tradeoff; moreover,the presence of skirts creates the low-frequency stop-band to suppressnear-field multiples 66 and creates the high-frequency stop-band toreduce imaging noise 68 in the extreme far field.

Of course, during any imaging, the gain of the receiver is increasedduring the echo time so that weaker deeper echoes are amplified morethan the stronger shallower echoes. This dynamically varying gain,presently currently used commercially is called "time gain compensation"(TGC). When applied along with the concept according to the presentinvention of a movable passband, the cumulative receiver performance canbe summarized in the bandpass curves 60A, 62A, and 64A of FIG. 10.

Additional advantages from the use of the sliding-frequency fixedbandwidth receiver mode arise in conjunction with the ultrasoundtransducer crystals. The ultrasound crystals designed to operate at alow frequency (e.g. 2.6 MHz) are capable of operating at higherfrequencies (e.g., 3.9 MHz) with reduced conversion efficiency. However,such a crystal is nevertheless capable of the higher azimuthalresolution associated with higher frequency processing. By employing acrystal cut to favor the low end of the frequency range (below thefrequency of the transmitted signal in CW systems), one is able tooptimize the receiver-crystal farfield performance and to minimizesubstantially the needed variation required in the TGC gain vs. timecontrol function. Being able to reduce TGC is important, because itreduces systems costs and minimizes intermodulation distortion in thereceiver's front end. This fact in itself is very important, becausefrequently a large (thick) reverberation in the body at moderate depthscontains sufficient low-frequency content to distort the receiver'sfunction so that thin, delicate anatomic line features are masked fromview due to intermodulation distortion when TGC reduces receiver gainsignificantly. In other words, by designing a signal-processing channelthat does not require as much TGC control, the overall signal-processingfidelity also improves with a given amount of hardware: alternatively,an excellent receiver (with fewer TGC-controlled stages) can be builtwith less hardware to attain reasonable image quality.

A further concept in variable-frequency ultrasound systems according tothe present invention is selecting a tone burst of a particular durationand selecting a tranceiver system impulse to produce a detected signalwhich is inherently deconvolved (the application of phase-synchronoustone burst). Looking at FIG. 11, the Active Gate Input (AGI) pulse 130is a kind of a master timer signal that initiates the onset of the pulse132, defines the echo time 134 and starts the delay gate time 136. Thetone burst 138 is a substitute test signal that simulates an idealizedsingle ultrasound echo. The duration 140 is adjustable, the tone-burstfrequency is adjustable, the delay gate 136 is adjustable; but the phaseof the start of the tone burst is always fixed with regard to the end ofthe delay gate time 136. Whenever the tone burst starts, 144, the finalreceiver (video) output 146 occurs. A long time burst 138 results in theconventional "unit step" video response. A short tone burst (e.g., 3cycles at 3.5 MHz) closely approximates a "squared-off" ultrasound echo.The tone burst 138, shortened to 3 cycles, causes the tail end 148 ofthe unit step response to move back in time until the two halves formthe axial impulse response 150. Demonstrating unit step responses thatshrink into axial impulse responses constitutes important tests tocharacterize ultrasound receiver functions. For instance, one normallydesigns the until step response to be overcompensated on a "squared-off"attack tone burst to end up with an "equalized" response to an actualonset attack coming from a crystal echo signal as summarized in FIG. 12.

This abbreviated discussion describes the acceptable transient responsefor any variable frequency tracking receiver designs. At first glance,one might expect an electrically tuned (varactor diode variety) singlesecond-order resonant circuit filter to be a satisfactory method ofachieving the movable bandpass of FIGS. 9 and 10. This is not the casebecause the tunable second-order filter has the wrong transientresponse, as illustrated in FIG. 13.

Referring to FIG. 13, a phase-synchronous tone burst 180 excites anenvelope 182 having a (1-e(t/t_(o))) multiplier when the bandpass isrestricted by a center-timed second-order resonant filter. Although therise time 184 can easily be scaled to render the proper axialresolution, the slope discontinuity 186 creates insurmountable(derivative-dependent) post-processing video enhancement problems. Evenwhen the second-order filter is mistuned, as a 190, theslope-discontinuity prevails (192), and the 0th-order video signal 194,196 is always slope-discontinuous. However, if a high-order (e.g.sixth-order) bandpass filter is used, the envelope response 200 isslope-continuous, and the resulting video 202 is also slope-continuous.In other words, an elegant (nth order) bandpass filter is preferredbecause it preconditions all receiver echoes in such a way that aslope-discontinuous video output is impossible, i.e., the signal canalways be deconvolved by a lead-function one derivative operator lowerthan the order of the effective bandpass filter function. The practicalvisually perceived counterparts of these principles are emphasized inFIG. 14.

The upper trace 210 of FIG. 14 illustrates a video signal originatingfrom an (improper) second-order bandpass filter. The axial extent of thepoint-spread function is elongated by the two (1-e(t/t_(o))) multipliers212, 214, and is generally smeared vertically (in time). The lower trace218 shows a much sharper axial imaging of the point spread function 220with the corresponding quicker attack 222 and decay 224 portions of thevideo waveform. The video performance of 218 can be obtained by the useof high-order filters having the same (10-90 percent) cumulative risetime as the single second-order filter because the slope continuity ofthe latter case permits post-video detection deconvolution to at leastthe first two derivatives. From a practical point of view, the bestaxial characteristic of a point spread function is a "blip" whose heightapproximates the desired axial resolution (about 1 mm) and contains veryslight (-10 percent) symmetrical shadow troughs above and below thecenter of the blip.

FIG. 15 illustrates a preferred system configuration for a baseband-RFmovable passband receiver. Since the objective of this receiver is tocorrect the signal processing for the most significant average effectthe body tissue has on the ultrasound pulse (summarized in FIG. 5), thename adopted for such receivers is "tissue signature trackingreceivers." A basic design of the receiver according to the presentinvention is shown in FIG. 15.

Referring to FIG. 15, the AGI timer 400 triggers the pulser 402 thatsends an avalanche breakdown pulse into thetransmit-receive/anti-transmit receive (TR/ATR) switch 404. This switch404 connects the probe 406 to the pulser during the pulse. During thetime that echoes are received by probe 406, the switch 404 disconnectspulser 402 and connects probe 406 to the input stage 408 of the receiver410. Meanwhile, AGI timer 400 triggers the ramp generators 412. Thefirst ramp function, the time-gain compensation (TGC) ramp 414 applies asaw-tooth shaped wave to gradually increase the gain of the preamplifierstage 408. The DC bias and slope of the TGC ramp are controllable bypanel potentiometers 416,418. At this point, RF echo signals produced atpoint 420 are infiltered and partially compensated in gain (the gain ispartially boosted to increase the amplitude of deep echoes), the signalat point 420 tends to have predominant low-frequency components (e.g. inthe 1.5-2.2 MHz range). By applying a time-adaptive high-pass filter422, the higher amplitude low-frequency components are attenuated beforethey can overmodulate the next RF amplifier stage 424. The TGC gaincontrol waveform is also applied to amplifier 424 to complete the job ofbalancing the near-and-far echoes to approximately the same signallevels. The variation in gain during echo time is about 20-25 dB up topoint 426, or about 10-13 dB per stage. The signal at point 426 istime-adaptively low-pass filtered at 428 in such a way that theremaining passband squeezing through filters 422 and 428 is of aconstant width and slides downward in frequency with time. The lower andupper tissue signature ramp generators 430 and 432 produce appropriatelyproportional waveforms to cause the filters 422 and 428 to trackproperly. The signal at point 434 is symmetrically logarithmicallycompressed at 436 to compress the tremendous variation (50-60 dB) inecho amplitudes into a smaller amplitude range (25-30 dB) more suitedfor envelope detection at 438 at reasonable (5-10 V pp) signal levels.Video detector 438 employs a second-harmonic carrier suppression filter(in the 4.4- to 7.0-MHz range) and a low-pass filter (700-1200 KHz) tolimit the rise time to one-half pixel height. Further enhancement anddeconvolution of the video signal is done at the deconvolver stage 440,also known as aperture correction. The analog video output 442 is fed tothe main display system; may be flash-digitized; entered into ascan-converter; or presented as an A-mode trace on an auxiliary monitor.

The waveform curves of FIG. 16 summarize how the various signals worktogether. The positive leading edge 500 of the AGI timing pulse 502initiates the outgoing acoustical pulse 504. Echoes 506 occurring duringecho time 508 have a generally decreasing amplitude vs. echo time. TheTGC ramp function 510 boosts gain during echo time, and then resets tothe lower gain value during the next pulse as controlled by the gain 512and slope 514 control settings. Meanwhile the lower TSG ramp function516 and 518 control the time-varying filters to create a constantbandwidth passband 520 that descends in frequency with echo time. Allfilter functions reset themselves to their initial values during eachsubsequent pulse. The width of the passband 522 determines the axialimaging resolution and the downward slope 524 determines how thereceiver tracks the decreasing frequency spectrum vs. echo depth due tothe tissue signature effect.

FIG. 17 provides another interpretation of how the two filters track intime. The cutoff 600 of the low-pass filter is actually higher than thecutoff frequency 602 of the high-pass filter by an amount equal to theeffective bandwidth. As the echo time progresses from 50 to 80 us afterthe pulse (progressing from near to far echoes) both filters 428, 422are constantly "reprogrammed" by ramp functions 432, 430 so that bothcutoff frequencies slide downward to preserve a constant differencebetween them 604. The constant bandwidth tracking filter design isaccomplished with a "partitioned" approach for several reasons. If asingle filter with movable bandpass were implemented, there would bedifficulty in changing the slope of the lower skirt without affectingthe upper skirt. A partitioned approach, employing two adjustablefilters with a buffer amplifier in between is much easier to design andto adjust at the testing stage. If one chooses to make the width of theeffective passband variable, the partitioned method very easily allowsfor this approach. If the shape of one or the other ramp function weremodified, this changes the bandwidth. There are imaging applications inwhich a reduced bandwidth at great depths in exchange for a bettersignal-to-noise ratio is the preferred mode. Conceivably, the offsetsbetween the two ramp functions could be at the disposal of the operatorwho could dial or program the tradeoff between axial (and azimuthal)resolution and dynamic range. An example of this approach occurs whenscanning the carotid arteries (in the neck). During the initial surveyof the scan, the operator might prefer a very wide bandwidth (high axialresolution, a sharp image) to identify the detailed architecture ofvessel walls (each branch of the carotid artery has four surfaces to beidentified). Once a suspicious plaque had been identified, the operatormight prefer to go into a reduced bandwidth mode (resulting in "softer"edges) in exchange for a higher signal-to-noise ratio and greaterdisplay dynamic range in order to classify the plaque and todifferentiate it among various stages of blood clot.

An additional feature of the present invention is the development ofsynchronous baseband RF detection. When one places a transducer againstthe human body and stops the scanning action, so as to repetitivelypulse-and-echo the same line of imaging information, an A-modepresentation of this information (similar to FIG. 6) has surprisinglygood phase-synchronous qualities. The ultrasound echo train will be manyoverlapping echoes. Each anatomical reflecting surface (or acousticalimpedance discontinuity) will produce echoes which are approximatelysingle-frequency CW packets inside of Gaussian multiplier envelopes. Theerror is that frequently the "CW wave inside the envelope" may not haveeither an absolutely "zeroed " baseline, shown by waveform 900 of FIG.18, or may not have a constant CW frequency, as shown by waveform 902.

The ultrasound imaging "pixel" is recoverable fully by envelopereconstruction (i.e. timing to the average frequency (function of depth)and perfectly detecting the (symmetrical) envelope). The nonzerobaseline defect 900 and the nonconstant CW frequency defect 902, ofcourse represent information, but for the next coming generation ofimaging refinements, can be eliminated to simplify signal processingsteps within the (narrowly defined) envelope reconstruction model. Themost practical method to eliminate defects 900 and 902 is to truncatethe receiver passband by employing fairly steep (6th to 8th order) skirtselectivity in the cumulative bandpass filter function prior todetection. When one forces the echo train to fit in the model of zerobaseline and constant CW frequency (per echo), the only dynamicallychanging feature of the entire echo train is that among hundreds ofechoes, only the CW frequency slowly slides downhill, parroting thegrossest body tissue signature physics (FIGS. 5 and 6). When one simplyenvelope-detects an echo train "laundered" of defects 900, 902, theresulting video information, although narrow banded, leads to imagesgenerally superior to totally conventional (broadband or TRF)processing. However, an echo train, laundered of defects 900 and 902,presents a wonderful opportunity to recover motion-related information,owing to its outstanding phase-synchronous characteristics.

However, according to the present invention, when one thinks of the"phase synchronous" concept in new light, significant improvements inultrasound transceivers are made, as discussed as follows. Assume that a"reference signal" was started or "kicked off" exactly at the same phaseangle every time AGI came up, as shown in FIG. 19, regardless of whethersuccessive AGI pulses occur at a regular repetition rate.

If the imaging signal processing were to function on a T₁ time basis,and forgetting about nonuniform "dead times" T₂, the system would act asthough it were totally phase-synchronous in the CW sense, even thoughthere is random phase breakup during nonconstant times T₂. The "phasesynchrony" during T₁ can easily be displayed on an oscilloscope simplyby triggering the scope of the positive-going edge 906 of AGI 908. It isimportant that an ultrasound signal processor be able to operate on anAGI-asynchronous basis (nonconstant T₂) because many commercial scannersdo and will operate on what is known as "demand pulsing" (i.e., waituntil the crystal beam is exactly where it is desired, and then initiatethe pulse (AGI)).

Although FIG. 19 describes a phase-synchronous gated fixed-frequencyreference signal, a much more useful concept is the phase-synchronousgated slow chirp sliding frequency reference signal. See FIG. 20. Thisreference wave 910 "kicks off" exactly synchronously with the positivegoing edge 912 of each AGI pulse 914. In addition, the reference wave910 slides downhill gradually in exactly the same way each time (910,918, 920) it gets "kicked off." It is useful to think of wave 910 as akind of "skeleton model" of the much more complex echo train.

According to one embodiment, the tranceiver of the present inventionuses the reference wave as a pilot tone to directly interact withincoming radio frequency (RF) signals at the baseband frequency. A pilottone is programmed to have a frequency-time modulation curve as shown inFIG. 21. The pilot tone frequency at times t_(a), t_(b), and t_(c)corresponds to the spectral peaks of acoustical reflections 40A, 42A,and 44A in sequence, of FIG. 22, from the human body coming back at thesame corresponding times.

According to the present invention, a pilot tone corresponding to theexpected CW component of any specific echo at any specific depth isgenerated. If the pilot tone only roughly approximates the CW frequencyof the echo, a very good reading of the "phase status" of that echo canbe taken because each echo contains very few (e.g. 3) cycles of radiofrequency. For instance, if the pilot tone 150 of FIG. 23 is "productdetected" (multiplied by the echo and LP-filtered) with the echo 152, anumber of output 154 (or no output) signals are possible.

Presentation of the pilot tone 940 only during a few microseconds makesit appear as a fixed frequency relative to the duration of the RF echosignal 942. If the two waves happen to be "in-phase," the productdetector output 944 results. If the pilot tone and signal are out ofphase, 946 results. If the pilot tone and signal are in quadrature, nooutput results. If the signal is represented as a phasor (see FIG. 24),the product detector outputs 944, 946, etc. can be thought of as thecosine projections of the envelope.

In FIG. 24, the echo phase 950 is shown not perfectly in phase with thepilot tone 952, so the cosine mechanics can be seen. To correspondexactly with the waves 940, 942 (of FIG. 23), one would have to rotatethe signal 954 towards the real axis 956. The product detector outputs(PDO) of FIG. 23 are like the conventional video signal with theseexceptions: (1) PDO may be bidirectional; (2) PDO may be zero event evenin the presence of a strong reflection; and (3) PDO "whips up and down"wildly for the slightest amount of motion (approximately 0.1 mm) of thereflecting anaotomy relative to the transducer crystal. Since for everycosine there is a corresponding sine, the null outputs 960 of FIG. 23can be remedied by duplicating the detection process. If the pilot tone940 is phase shifted by 90° for all frequencies used during the "chirp"of the pilot signal, one could employ both the original andphase-shifted pilot tone to operate two product detectors to produceboth the sine and cosine component outputs as suggested in FIG. 25.

The phase angle of the echo signal 970 is unpredictable. By making areceiver for which the phase angle is irrelevant, one can gather boththe amplitude and phase information contained within each pixel withoutrequiring the receiver to identify zero crossings in signal or worryabout locking any oscillator into an incoming carrier. Therefore, thereceiver according to the present invention is a "nonthresholding"receiver, which performs smoothly as the received signal drops into thenoise level of the receiver and received noise signals. The presentinvention therefore provides information for diagnostic imaging even asthe signal-to-noise (S/N) ratio drops below 0 dB. The outputs of thisreceiver are the two product detector outputs, and both conventional(unidirectional amplitude) video and motion-related imaging informationcan be derived from these two outputs.

FIG. 26 shows how to implement this receiver. The front end blocks 1000,1002, 1004, 1006, 1008, 1014, 1022, 1030, 1024, 1032, 1028, and 1036 ofFIG. 26 work in the same way that blocks 400, 402, 404, 406, 408, 422,430, 424, 432, 428, and 436 of FIG. 15 do. An additional output 1060 ofthe AGI timer 1000 feeds into the gated-start VCO 1062 to "kick it off"in the correct phase when AGI comes up. An extra "pilot tone" rampgenerator is provided at 1064, whose purpose is to provide a sawtoothwaveform at an AGI rate (similar to the TGC and TSC ramp functions) thattunes a varactor-diode operated resonant tank circuit in the pilot-toneVCO 1062. The output from the VCO 1062 is applied directly to multiplier1070 and through phase shifter 1072 to multiplier 1074. Signals fromlogarithmic compressor 1036 are also applied to multipliers 1070 and1074. The outputs 1076 and 1078 are lowpass filtered at 1080 and 1082and deconvolved at 1084 and 1086 in the conventional manner. The mosteconomical location for the log compressor 1036 is before themultipliers 1070 and 1074 provided that one takes the precaution toreject the fundamental RF component at lowpass filters 1080, 1082. Themathematics for recovering the sine and cosine components in theircorrect ratios works with the log compressor before the multiplier andall outputs 1088 and 1090 are equally logarithmically compressed. Thealternative design (i.e. using two log compressors after multipliers1070 and 1074) is more expensive and incurs the problem that themultipliers have to handle 60 dB of dynamic range rather than 30 dBrequired if the single log compressor is used ahead of both multipliers.In order to recover both amplitude and velocity related information,outputs 1088, 1090 must be post-processed. One preferred method,outlined in FIG. 27, is to digitize both outputs and do the computationsthrough EPROM look-up tables 1118 and 1120.

Referring to the receiver post-processor shown in FIG. 27, outputs 1088,1090 are coupled through opto-isolators 1100, 1102 before beingflash-digitized at ADCs 1104, 1106. Routine digital timing 1108, strobes1110, and incidental logic 1112 is condensed for simplicity in the mainsignal path. Digital code equivalents 1114, 1116 of the real andimaginary signals are combined so that the sum of all the bitsrepresents the address code inputs for EPROMs 1118, 1120. For everypossible combination of real and imaginary echo signal components, EPROM1118 digitally recalls the code value for the "square root of the sum ofthe squares" and provides a "Digital Video Output" (DVO) 1124, which isthen converted back to analog at DAC 1122 and presented as "Analog VideoOutput" (AVO) 1126. Meanwhile, the digital codes 1114, 1116 combine toform collective addresses to the arctan ERROM 1120 whose output 1128 isthe digital equivalent of the arc-tangent of the real and imaginarysignals. Since there is no phase alignment between the chirped pilottone and RF echo signals, the train of arc-tangent values is a randomcollection of angular numbers for any single echo train (line of imagingdata). When any echoes move from one AGI time to the next, thearc-tangent values only for those moving echoes change values, and allother values remain the same. In order to detect such motion, on apixel-by-pixel basis, the entire train of hundreds of pixel-sizearc-tangents is stored in the one-line static RAM 1130 (an N-bitsubstitute) for a shift register, capable of variable-length AGI on-timeand is played back with a one-AGI-cycle delay. The delayed playback RAMoutput 1132 is subtracted digitally 1134 from the next line ofpixel-sized arc-tangents, and the output 1136 is a digital equivalent ofa bidirectional velocity signal listing all velocity motionscontinuously down the echo time as a direct motion counterpart of theordinary A-mode amplitude signal.

In order to determine which velocity data is to be utilized by thedisplay system, a thresholder 1138 with High 1140 and Low 1142 controlsis provided. FIG. 28 illustrates its echoes in the 0- to -30-dB range1120 (after equalizing with the TGC camps) represent anatomicalfeatures, while echoes in the -30- to -55-dB range 1202 represent bloodflow. If the thresholder controls 1140, 1142 of FIG. 27 are set to 0 dBand -30 dB respectively, digital velocity data 1136 will pass throughthe digital muting circuit 1144 to allow the "Digital Doppler Output"(DDO) 1146 to be used to display anatomical motion (such as minutepulsations of arteries). DAC 1148 provides an "Analog Doppler Output"(ADO) 1150 so that the velocity signal can be observed along with theamplitude signal on a scope for diagnostic system check purposes.

If the thresholder controls 1140 and 1142 are set to -30 dB and -55 dBrespectively, digital velocity data 1136 will pass through the mutingcircuit 1144 only for very weak echoes that are normally black in theconventional ultrasound impage (0 dB=white, -15 dB=grey, -30 dB=black).The lower thresholder settings (-30, -55 dB) are very useful becausethey allow the main system imaging this data to differentiate betweendark vessels in the image and cystic voids that have loculated(nonflowing) liquid from dark regions having blood flow.

Naturally, the velocity signal is proportional to the magnitude (incm/sec) of the flow, is bidirectionally sensitive, is proportional tothe cosine of the angle between the ultrasound beam and the blood vessel(or anatomical motion direction), and is proportional to the pilot tonefrequency.

Additional features of the present invention include:

(1) Two or more scanning probes might be used to cover the case whereinone probe is exactly perpendicular to the (site and) direction of motionand suffers a complete null output;

(2) A microprocessor based system that "expands and compresses" thevelocity numbers to include the 1/cosine θ compensation factor betweenthe angle of the beam and the long axis of the blood vessel beingimaged;

(3) Digital correction proportional to 1/f_(PT) of the pilot tonefrequency, to normalize velocity data to a uniform scaling factor alongthe changing frequencies during the entire echo time;

(4) Digital correction that corrects for different rates of change inthe arc tangents due to changes in the pulse repetition (T₂) rates ofthe ACI master timing pulse. One way of minimizing the Doppler aliasingproblem is to allow the AGI timer to be agile, i.e., always pulse at themaximum possible rate consistent with the penetration depth selected.The processor described in FIG. 27 is designed for "low clutter"signals. FIG. 29 shows what is meant.

The echo phasor hovers and rotates quite wildly, depending upon themotion involved, but is always large enough compared to any interferingsignal that it describes a doughnut about the origin of the graph. Ifthe rate of rotation 1230 of the arc tangent is "jittery," it can stillbe used provided that the average cyclic rotation (one-turn rate) ispreserved.

FIG. 30 shows what is meant by a high-clutter echo. In this case, theecho phasor 1240 is always somewhere in the "Christmas wreath" region1242 and exhibits an oscillating nature (envisioned as the drive-rodmotion on a steam locomotive). This representation occurs because theecho of interest is smaller than a large (generally stationary)interfering clutter signal (such as a body "multiple" from an earlierimaging feature). In order to retrieve true velocity information fromthis offset form of circular motion, a digital processor can translatethe origin as shown in FIG. 31 which operates as follows.

The echo phasor positions must be stored for several contiguous echoes(same pixel), the average position 1250 computed, and the phasors musthave the average value subtracted in order to affect the equivalent of acoordinate translation. Subsequent subtraction of corrected phasors willrestore some velocity information; the quality of this informationdepends upon how stationary the clutter signal phase happens to be.Remember, the "clutter signal" may not really be clutter, but adifferent category of a velocity signal in itself. From an imaging pointof view, it only takes two contiguous (adjacent) imaging lines in areal-time image to recover low-clutter Doppler motion information (e.g.,showing blood flow in major vessels). Since scanners normally spacedisplay line closer than the azimuthal resolution distance of thecrystal, a 2:1 reduction in motion-azimuthal resolution is hardly adegradation over anatomical imaging. However, motion recovery in thepresence of high clutter requires several lines of computaton, and thisdegrades motion-azimuthal resolution by 5:1 typically.

If one were to image anatomy in monochrome (as is done commercially), avery convenient way to image motion in a composite fashion would be toraise the image "in relief"proportional to the flow values. This wouldgive the image a three-dimensional effect where flow is significant.Another way would be to use a series of closely spaced "highlighter"lines in the display that bend to and fro like spider webs in thebreeze. In this manner, the familiar parabolic flow profile in an arterycould be seen directly.

Another method of composite display is to image the anatomy inmonochrome and the motion in color. Suitable color scales can be arrivedat that have simple interpretations; e.g., reddish shades couldsymbolize blood flowing towards the crystal; blue symbolizes flow away.It is important that whatever color scale is used, that the maximum flowvalue color away from the probe chromatically connects with the maximumflow valve towards the probe, because if this is done, aliases becomeinsignificant distortions in the imaging process. It is also importantthat the color scale selected goes to black, so that the polychromaticimaging of cysts remains black, as it is currently in commercialmonochrome, for zero-velocity conditions.

Color imaging of motion can provide impressive interpretive results,based on what we already know about color television engineering. Theeye is quite content to see less color detail (axial resolution,bandwidth, and azimuthal resolution) than monochrome detail. Inchromatic ultrasound, it is the monochromatic (square root of the sum ofthe square) part that does all of the "work" of producing the image: the(reduced bandwidth-permissible) color velocity component is an aid thatcategorizes the anatomic feature. The color detail can be aboutone-fourth as dense as the monochrome detail before the eye discerns theloss of information. In the event of high clutter contamination, forinstance, a moderate amount of reduction in azimuthal resolution is welltolerated.

It should be noted that the use of processors of FIGS. 26 and 27 treatthe axial resolution component of velocity with the same bandwidth(pixel count per line) as the axial resolution component of echoamplitude. An alternative processor design might be to split thebandlimiting section 1009 (of FIG. 26) into two sections; a wider bandsection for a conventional receiver of high axial resolution and anarrower band section for the lower axial resolution Doppler processing.This alternative design has the advantage of preserving velocitydetection over a larger dynamic range (with less spatial resolution),but has the disadvantages of bulk and cost.

Other embodiments and modifications of the present invention by oneskilled in the art are within the scope of the present invention, whichis not to be limited, except by the claims which follows.

What is claimed is:
 1. For use with an ultrasound transducer, a selectedbandwidth ultrasound tranceiver comprising:means for generating anultrasound signal having a selected amplitude and frequency; means forenergizing said transducer for signal propagation through a mediumaccording to said ultrasound signal, wherein said transducer provides areflected signal upon recipt of a reflection of said propagated signalwith said medium; and means for receiving said reflected signal,comprising:filter means receiving said reflected signal and havingselectively decreasing high and low frequency cutoff characteristics forincreasing depth of reflections of said propagated signal; and detectormeans providing an output signal according to the output of said filter,wherein said detector means output signal describes the reflectivecharacteristics of said medium, the amplitude variations and relativetime intervals provide indication of medium reflectivity atcorresponding medium depth.
 2. The selected bandwidth ultrasoundtranceiver of claim 1, whereinsaid filter means comprises a partitionedtracking filter having a highpass filter and a lowpass filter, whereineach said highpass and said lowpass filter is selected to form aselected bandpass, the filter characteristics of each said highpass andlowpass filter being independently controlled to pass the desired signalfrequency components.
 3. The selected bandwidth ultrasound tranceiver ofclaim 1, whereinsaid filter means provides a constant bandwidththroughout the range of operation.
 4. A method of processing imagingsignals comprising the steps of:emitting a signal burst of a selectedduration into a medium wherein the image-related informationcharacteristics are degraded according to the medium characteristics;receiving energy from said medium according to said signal burstproducing a received signal therefrom; filtering said received signalaccording to a filter having selectively adjusted high and low frequencyresponse cutoff characteristics with said cutoff characteristics beingadjusted in a same direction and detecting said filtered receivedsignal, wherein filter response characteristic and said signal burstduration are selected to provide a detected filtered signal, said filterhaving a characteristic inverse to said medium related imagedegradation, to compensate for said degraded image characteristic. 5.The method of claim 4, whereinsaid filter response characteristic is atime-continuous function.
 6. The method of claim 5, whereinsaid toneburst duration is selected to produce a detected filtered signal havinga deconvolution impulse response characteristic.
 7. The method of claim6, further including the step ofdisplaying an image according to saiddetected filtered received signal.
 8. The method of claim 7, whereinthemedium is an acoustic medium having image-degrading characteristics. 9.An ultrasound tranceiver, comprising:transmitter means for producing atone burst of selected duration into an acoustic medium having animage-degrading characteristic; receiver means for producing a receivedsignal according to the reflected energy from said tone burst in saidmedium; means for filtering said received signal producing an outputsignal having a selected response characteristics; said filtering meanshaving selectively adjusted high and low frequency response cutoffcharacteristics with said cutoff characteristics being adjusted in asame direction; and means for detecting said filter output signal,wherein at least one of a filter response characteristic and said toneburst duration are selected to provide a detected filter output signalhaving a characteristic inverse to said image-degrading characteristicsto compensate for said acoustic medium image degrading characteristic.10. The ultrasound transceiver of claim 9, whereinsaid filter responsecharacteristic in a time-continuous function.
 11. The ultrasoundtranceiver of claim 10, whereinsaid tone burst characteristic isselected to produce a detected filter output signal having adeconvolution impulse response characteristic.
 12. The ultrasoundtranceiver of claim 11, further includingmeans for displaying an imageaccording to said detected filter output signal.